2008 IEEE INTERNATIONAL RF AND MICROWAVE CONFERENCE PROCEEDINGS 2008 IEEE INTERNATIONAL RF AND MICROWAVE CONFERENCE PROCEEDINGS 2008 IEEE INTERNATIONAL RF AND MICROWAVE CONFERENCE PROCEEDINGS 2008 IEEE INTERNATIONAL RF AND MICROWAVE CONFERENCE PROCEEDINGS December 2December 2December 2December 2----4, 2008, Kuala Lumpur, MALAYSIA4, 2008, Kuala Lumpur, MALAYSIA4, 2008, Kuala Lumpur, MALAYSIA4, 2008, Kuala Lumpur, MALAYSIA

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Code Rate Adaptation for Mitigating the Effects of Detect And Avoid (DAA)

Mechanism on the Performance of UWB Systems

A. M. Rateb, S. K. Yusof and N. Fisal

Faculty of Electrical Engineering (FKE), Telecommunication Research Group (TRG),

Universiti Teknologi Malaysia (UTM), Skudai, Johor, Malaysia.

ahmadmrateb@ieee.org, kamilah@fke.utm.my, sheila@fke.utm.my.

Abstract – Ultra wideband (UWB) systems can cause

potential interference to other narrowband systems

sharing the same band, such as WiMax and Wi-Fi.

Detect And Avoid (DAA) mechanisms were suggested

as a cognitive radio-based solution to reduce

interference by adaptively reducing the transmitted

power at the overlapping bands. Applying this

mechanism may result in degrading the performance of

UWB transmissions. In this paper, we suggest a minor

modification to the DAA mechanism to mitigate this

effect while preserving data rate, and thus providing

UWB with more reliability. Simulation results show

significant improvement in performance over various

interference scenarios and data rates.

Keywords: Detect and Avoid (DAA); Cognitive radio; Ultra

wideband (UWB); MB-OFDM; Spectrum shaping

1. Introduction

Since the year 2002, there has been a growing

interest in UWB systems due to the large band (3.1 to

10.6 GHz) allocated by the FCC for these systems.

UWB is planned to work as a spectral underlay system

for short-range, low-power communications [1]. Being

underlay, UWB is required to coexist with existing and

potential high power narrowband systems sharing its

spectrum without causing any unacceptable

interference to them. Examples of existing systems are

IEEE 802.11a systems (4.9-5.85 GHz) and WiMax

systems (3.3-3.8, 5.4-5.825 GHz). This led the FCC to

require UWB transmissions to adhere to a spectral

mask limiting its transmitted power to -41dBm. In

spite of this very low power constraint, interference

between UWB and narrowband systems can still be

significant if both devices were present in a close

proximity.

Cognitive radio (CR) technology [2] offers novel

methods for improving spectrum efficiency and

transmission reliability of communication systems. It

adds spectrum awareness feature to communication

systems to attain better adaptation to different channel

conditions, and exploit available spectrum

opportunities through dynamically changing

transmission parameters. Many recent studies have

focused on finding CR-based solutions to allow

coexistence between UWB and narrowband systems.

The work in this area was combined and formulated

into what is called “Detect And Avoid (DAA)”

mechanism [3], designed for UWB systems based on

Multiband Orthogonal Frequency Division

Multiplexing (MB-OFDM) technology [4]. This

mechanism mainly focuses on continuous detection of

narrowband transmissions and avoiding them by

selectively reducing the transmitted power in the

overlapping victim band. This Reduction is usually

accompanied by loss of UWB user data carried on this

band. Recovering this data is left to the redundancy

added to the transmitted data; however, if the

overlapping band was large, or the added redundancy

was not sufficient, data transmission might not be

successful [3].

In this paper, we introduce a modification to the

current DAA mechanism that acts to mitigate its

effects on UWB link performance. The modification is

mainly based on preventing data loss by dumping

transmission of data on the victim band, and

redistributing this data on the remaining band. Data

rate throughput is sustained by adapting the channel

coding rate to handle the added data payload.

As for the remaining part of this paper, we provide

an overview over the PHY layer of UWB systems in

section 2. In section 3, we briefly introduce the DAA

mechanism. In section 4, we explain the modification

we propose, its operation, and performance. We

present our results in section 5, and draw our

conclusions in section 6.

2. UWB PHY layer

In this section, we provide a short overview over

the relevant features of MB-OFDM UWB systems as

pointed out in the ECMA-368 standard issued by the

WiMedia alliance for high rate applications [5].

In MB-OFDM approach, the available spectrum

of 7.5 GHz is divided into 14 bands of 528 MHz each.

The 14 bands form 5 band groups, where each band

group contains 3 bands except for the last which

contains two. For transmission on one of these band

groups, the standard employs a frequency-hopping

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technique in which the carrier frequency is hopped

between the center frequencies of its bands after each

OFDM symbol. This is done according to a predefined

Time-Frequency Code (TFC). Unique logical channels

that can be used simultaneously by multiple UWB

piconets are defined by using different TFCs for over

each band group. The first band group (3.17-4.75

GHz) has been decided for operation of first

generation UWB devices.

MB-OFDM transceiver is based on a conventional

OFDM transceiver that uses a total of 128 OFDM

subcarriers divided into 100 data subcarriers, 12 pilot

subcarriers, 10 guard subcarriers, and 6 unused

subcarriers. For low data rates, MB-OFDM system

uses Quadrature Phase Shift Keying (QPSK)

modulation; whereas for high rates, it uses Dual

Carrier Modulation (DCM) to increase diversity at the

expense of receiver complexity. DCM introduces

additional redundancy by mapping 4 bits onto two 16-

point constellations. The symbols from the two 16-

point constellations are then mapped onto tones that

are separated by at least 200 MHz in bandwidth.

The standard provides a set of four convolutional

code rates which are: 1/3, 1/2, 5/8 and 3/4. The last

three code rates are generated from the mother rate 1/3

by employing code puncturing. For improved

performance over fading channel conditions, the MB-

OFDM standard supports Frequency Domain

Spreading (FDS) and Time Domain Spreading (TDS)

for low data rates. In FDS, the same data symbol is

transmitted twice over two widely separated

subcarriers in the same OFDM symbol; whereas in

TDS, the same OFDM symbol is transmitted twice on

two consecutive time slots.

The standard supports avoidance of other

narrowband users allocated within the UWB band in

order comply with potential DAA requirements; the

transmitted signal is sent in the context of a Tone

Nulling array (TN) of 384 elements. These correspond

to the total number of subcarriers of the band group

used for transmission; so that TN [0 to 127] apply to

the subcarriers of the lowest frequency band in the

current band group, TN [128 to 255] to the middle

band, and TN[256 to 383] to the highest band. Each

tone nulling element can take the values of 0 or 1; if

the value is 0, then the transmitter should take steps to

minimize the transmitted signal power at the frequency

of the corresponding subcarrier; if the value is 1, then

the signal is unaffected by tone nulling. Tone nulling

applies to all symbols, including the preamble

sequence.

3. Detect And Avoid (DAA)

In a DAA mechanism, before the UWB

transmitter attempts to transmit data on a certain band,

it first scans this band to detect any surrounding active

narrowband links within this band, and measures their

power levels. Measured power levels for detected links

that require an avoid action should exceed a certain

threshold that indicates possible interference hazard.

The bands of the detected links are mapped onto the

TN array by setting corresponding array elements to 0.

Avoid acts to protect the narrowband link by ceasing or

reducing transmit power on the victim band to create a

notch in the transmitted signal spectrum at this band.

This implies that the data carried on the notched

portion of the transmit spectrum is spoilt since the

Signal to Noise plus Interference Ratio (SNIR) at

victim band will be exceedingly low. Redundancy

provided by FDS, TDS, and channel coding in low

data rates may be exploited to recover this data. In [3],

the target Packet Error Rate (PER) of 8% required by

the standard could be reached for low data rates (with

lowered link margins). However, for high data rates,

the target PER could not be reached for any transmit

power.

4. Code rate adaptation

The WiMedia standard specifies ten different

TFC’s for each band group (except for Band Group 5).

Each TFC defines a different frequency hopping

pattern that may hop between three or two bands, or

transmit on one band without hopping. Consequently,

the size of the TN array generated at the detect stage in

the DAA mechanism will vary according to the

number of bands employed in the TFC. The TN array

will contain 128 or 256 or 384 elements if the TFC

employs one or two or three bands, respectively.

Within the TN array (no matter its size), let the detect

stage has generated a total of ∆ elements that are set to

‘0’ requiring corresponding tones to be nulled. ∆ may

correspond to a mixture of data, pilot, and guard

subcarriers. If δ data tones are nulled within ∆, they

can be extracted by knowing the tone mapping

specified by the standard.

Each data rate uses a certain channel coding rate

C. Assuming δ data tones required to be nulled, we

shall cease transmission of data on these tones;

however, we will use the remaining data tones to

transmit the same number of information bits by

incrementing the channel coding rate (less coded bits

for the same number of information bits) with an

appropriate factor. For 100 data tones per band, the

required code rate is

100

'100

C Cλ

λ δ= ×

− (1)

where λ depends on the TFC used, and equals to the

number of bands the transmitter hops between; hence,

λ can have the values of 1, 2 and 3. Equation (1) shows

that C’ can take various values according to different

values of δ. This requires a flexibility of code rates that

cannot be accomplished using regular coding schemes.

To achieve the required code rates, we designed a

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scheme that can provide us with the code rates needed;

we introduce this scheme below.

4.1 Hybrid rate (HR) channel coding scheme For a generic channel encoder, a block of K

uncoded information bits is input to the encoder, and

output as n coded bits. In convolutional coding, n is an

integer multiple of K. The output of the encoder can be

punctured to achieve more code rates.

Suppose that the input K-bits block is divided into

M sub-blocks of sizes k1, k2...kM bits, then each sub-

block is fed into a separate convolutional encoder

amongst a set of M convolutional encoders with code

rates C1,C2...CM. The output blocks of size n1,n2...nM

bits are then concatenated to form the original data

block of n bits. We can achieve an overall code rate

C’=K/n as required in (1) if the following two

conditions are satisfied

1

M

i

i

k K=

=∑ (2)

1 1

M Mi

i

i ii

kn n

C= =

= =∑ ∑ (3)

it can be noticed that, to satisfy conditions (2) and (3),

at least one rate among the M rates should be higher

than the required code rate, and one is lower.

For simplicity, we chose M=2 to limit the number

of encoders/decoders needed to two. To eliminate the

need for modifying the original encoder, we chose the

first rate CL to be the next lower rate than C’ in the

standard code rate set, and the second rate CH to be the

next higher rate. Figure 1 demonstrates how HR

coding operates; the K bits input block is divided into

two sub-blocks of k1, k2 bits. The k1 bits are input to

the encoder with rate CL, while the k2 bits are input to

encoder of rate CH. The two output blocks of n1 and n2

bits are then concatenated to form one block of n bits,

and fed to the interleaver.

Code rate adaptation compensates for the reduced

number of data subcarriers by incrementing the code

rate by one increment for each nulled tone, starting at

the minimum available rate Cmin until it reaches the

maximum allowable rate Cmax. Cmin is the original code

rate C of the current data rate; on the other hand, Cmax

is specified according to performance requirements. In

this paper, we assume that the minimal performance

requirements are satisfied by setting Cmax to the highest

code rate employed by the standard, which is 3/4. This

implies that the performance of the data rate 480Mb/s

cannot be improved because as shown in Table 1, it

already employs code rate 3/4; however, it can if

higher value of Cmax was used.

The maximum number of data subcarriers that can

be disabled δmax can be derived from (1) by

substituting C with Cmin and C’ with Cmax to yield

min

max

max

100 1C

Cδ λ

= −

(4)

such that x denotes the largest integer less than or

equal to x. δmax values for all data rates are calculated

and listed in Table 1. In order to optimize

performance, CL and CH values are chosen for each

required code rate such that the highest possible value

for CL, and the lowest possible value for CH are used. CL and CH values for possible ranges of δ for data rates

with the same Cmin are given in Table 2.

Assume that a block of N coded bits is transmitted

over J OFDM symbols. J is required to be a multiple

of 6 since TFC patterns repeat every 6 OFDM

symbols. Assuming that the TN array will remain

unchanged during the J symbols duration, if data tones

are required to be nulled, code rate adaptation reduces

the N bits to n bits according to the number of nulled

data tones. n can be expressed as

2J

n NS

δλ

= − (5)

where S is the spreading factor, and equals 4 if both

TDS and FDS are used, equals 2 if either TDS or FDS

are used, and equals 1 if neither are used. The factor

‘2’ is the average number of bits per symbol. For

simplicity, we will not consider frequency hopped

operation for the rest of this paper; we will assume that

transmission takes place on one band; hence, λ is set to

1 in (4) and (5), and the TN array is limited to 128

elements. By combining (2) and (3), k1 can be

expressed as

( )1 Hk C n Kγ= − (6)

such that γ =CL/(CH−CL), it follows that k2 = K−k1. We

found that for k1 and k2 to have integer values for all

possible number of nulled data tones at all data rates,

N has to be a multiple of 1200 bits. K, N values for an

example block consisting of 24 OFDM symbols are

listed in Table 1 for all data rates.

Figure 1: HR coding for code rates between CL, and CH.

Decoding at the receiver is the opposite operation

to the encoding process; input coded blocks of n bits

are split into two sub-blocks of size n1 and n2 and fed

into decoders with rates CL and CH, respectively.

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Output blocks of size k1 and k2 are then concatenated

to form the original data block of size K bits.

Hardware requirement for HR coding is mainly to

add one duplicate of the convolutional

encoder/decoder and puncturer/de-puncturer used by

the standard to work in parallel with the original ones.

This signifies no added encoding or decoding

complexity since both operations are done in parallel.

Table 1: Parameters related to standard data rates.

Data rate

Mb/s

FDS

TDS

Cmin

Cmax

δmax

N / 24 OFDM

Symb.

K / 24 OFDM

Symb.

53.3 YES YES 1/3 55 1200 400

80 YES YES 1/2 33 1200 600

106.7 NO YES 1/3 55 2400 800

160 NO YES 1/2 33 2400 1200

200 NO YES 5/8 16 2400 1500

320 NO NO 1/2 33 4800 2400

400 NO NO 5/8 16 4800 3000

480 NO NO 3/4

3/4

0 4800 3600

4.2 Operation at the transmitter and receiver Code rate adaptation at the transmitter examines

the data rate and the required number of disabled data

subcarriers. From the given data rate, the transmitter

determines the maximum allowed number of nulled

data tones (δmax). In case the required number is larger

than δmax, data rate is forced to be reduced to the next

lower rate; if it is not, then the transmitter proceeds

and calculates the parameters k1, k2, CL and CH, and

configures the puncturers and encoders in accordance.

Then, it extracts sub-blocks of size k1 and k2 from the

input data stream and inputs them to the encoders.

At the receiver side, the TN array must be

available to successfully decode received data. The

receiver could be informed by the transmitter, or could

blindly generate the TN array based on its own

measurements. Since this issue is outside the scope of

this paper, we assume the receiver to be aware of

changes in the transmission parameters and that the

TN array is already available. By analyzing the TN

array, and the current data rate, the receiver indentifies

δ, calculates the parameters CL, CH, n1 and n2, and

configures the de-puncturers and decoders. Then, it

extracts sub-blocks of size n1 and n2 from the input

coded data stream and inputs them to the decoders.

4.3 Performance of HR coding scheme Given that data is distributed over M encoders

with M different code rates, resulting in M output sub-

blocks of n1, n2…nM bits. Assuming that the coded bits

from all sub-blocks are well interleaved to distribute

error events uniformly over all sub-blocks, the overall

probability of error can be written as

1

1 M

i i

i

P n Pn =

= ∑ (7)

such that Pi is the probability of error for data encoded

by rate Ci convolutional code. For M =2, (7) reduces to

( )1 1 2 2

1P n P n P

n= + (8)

where P1 and P2 are the error probabilities

corresponding to CL and CH, respectively. Equation (8)

indicates that the performance of any code rate

between CL and CH is lower bounded by the

performance of code rate CL , and upper bounded by

the performance of code rate CH.

Table 2: Parameters related to HR coding scheme.

Cmin CL CH γ δ

1/3 1/2 2 0 - 33

1/2 5/8 4 34 - 46

1/3

5/8 3/4 5 47 - 55

1/2 5/8 4 1 - 20 1/2

5/8 3/4 5 21 - 33

5/8 5/8 3/4 5 0 - 16

5. Simulation results

To evaluate and compare the performance of our

proposed modification to that of the original DAA

mechanism, we carried out BER simulations for UWB

transmission at various numbers of nulled data tones,

with and without our proposed modification. The

simulations were performed in non-frequency hopped

mode with soft decision Viterbi decoding. Since FDS

and TDS are mainly added to improve performance on

fading channels, we chose to perform the simulations

without including TDS or FDS, and on AWGN

channel to be able to monitor the direct impact of our

proposed coding scheme on error performance. This

implies that data rates having the same code rate will

perform similarly. The simulations were done

assuming QPSK modulation for all data rates

including those using DCM, based on the fact that

DCM performance over AWGN channel is equivalent

to that of QPSK [6].

Figures 2(a), 2(b) and 2(c) demonstrate the BER

performance plots for data rates with original code

rates of 1/3, 1/2 and 5/8, respectively. The plots

compare the performance of original DAA mechanism

without modification to the performance of our

modified version. It is obvious in these figures that

applying our modification improves performance in all

cases; whereas improvement is most significant for

larger numbers of nulled tones, since the degradation

resulting from nulling more tones in case of code rate

adaptation is very small compared to the case where no

action is taken to prevent the loss of data carried on the

nulled tones. Increasing transmitted power on the

99

remaining non-nulled tones provides minimal

improvement to the original DAA mechanism

performance, since the data lost due to tone nulling is

the major source of errors, while performance further

improves in our modified version since no data is

carried on the nulled tones.

6. Conclusions

In this paper, we proposed a modification to the

current DAA mechanism that mitigates its effects on

the performance of UWB systems, and we introduced

a channel coding scheme that can achieve high code

rate flexibility without increasing decoding

complexity. Simulated results show significant

improvement of performance upon applying our

modification. This can considerably reduce the PER

without reduction in link margins, even for high data

rates. This methodology can be applied not only to

UWB, but also to any OFDM-based system, either

underlay or overlay, undergoing spectrum shaping.

References

[1] Federal Communications Commission (FCC).

Revision of Part 15 of the Commissions Rules

Regarding Ultra-Wideband Transmission

Systems. First Report and Order, ET Docket 98-

153, FCC 02-48; Adopted: February 14, 2002;

Released: April 22, 2002.

[2] S. Haykin, "Cognitive radio: brain-empowered

wireless communications," IEEE Journal on

Selected Areas in Communications, vol. 23, no. 2,

pp. 201-220, Feb. 2005.

[3] S. M. Mishra, R. W. Brodersen, S. T. Brink, and

R. Mahadevappa, "Detect and avoid: an ultra-

wideband/WiMax coexistence mechanism, " IEEE

Communications Magazine, vol. 45, no. 6, pp. 68-

75, Jun. 2007.

[4] Anuj Batra et al. Multiband OFDM Physical

Layer Proposal for IEEE 802.15 Task Group 3a.

IEEE P802.15-03/268r3, March 2004.

[5] High Rate Ultra Wideband PHY and MAC

Standard, ECMA International ECMA-368, 2nd

edition, Dec. 2007.

[6] P. Ki-Hong, S. Hyung-Ki, and K. Young-Chai,

"BER Analysis of Dual Carrier Modulation Based

on ML Decoding,” in Proc. of Asia-Pacific

Conference on Communications, 2006. APCC '06.

, Aug. 2006, pp. 1-4.

(a)

(b)

(c)

Figure 2: Simulated BER performance over AWGN

channel for various numbers of nulled data tones. (a)

Cmin=1/3; rates: 53.3, 106.7 Mb/s. (b) Cmin=1/2; rates: 80,

160, 320 Mb/s. (c) Cmin=5/8; rates: 200, 400 Mb/s.

100